Probabilistic shaping techniques for high performance coherent optical transceivers

ABSTRACT

A method and structure for probabilistic shaping and compensation techniques in coherent optical receivers. According to an example, the present invention provides a method and structure for an implementation of distribution matcher encoders and decoders for probabilistic shaping applications. The techniques involved avoid the traditional implementations based on arithmetic coding, which requires intensive multiplication functions. Furthermore, these probabilistic shaping techniques can be used in combination with LDPC codes through reverse concatenation techniques.

CROSS-REFERENCES TO RELATED APPLICATIONS

The present application claims priority to and is a continuation of U.S. Application Ser. No. 17/077,755 filed on Oct. 22, 2020, which claims priority to and is a continuation of U.S. application Ser. No. 16/797,704 filed on Feb. 21, 2020, which is a continuation of U.S. application Ser. No. 16/256,971 filed on Jan. 24, 2019, now U.S. Pat. No. 10,608,749, which claims priority to U.S. Provisional App. No. 62/621,521, filed on Jan. 24, 2018, all of which are incorporated by reference for all purposes.

BACKGROUND OF THE INVENTION

The present invention relates to communication systems and integrated circuit (IC) devices. More particularly, the present invention provides for improved methods and devices for optical communication.

Over the last few decades, the use of communication networks exploded. In the early days Internet, popular applications were limited to emails, bulletin board, and mostly informational and text-based web page surfing, and the amount of data transferred was usually relatively small. Today, Internet and mobile applications demand a huge amount of bandwidth for transferring photo, video, music, and other multimedia files. For example, a social network like Facebook processes more than 500 TB of data daily. With such high demands on data and data transfer, existing data communication systems need to be improved to address these needs.

Optical communication is one major technological area that is growing to address these high demands on data. Optical communication systems typically communicate data over a plurality of channels corresponding to different phases and/or polarizations of the optical signal. While the data communicated over the different channels is typically aligned relative to a common clock when transmitted by the transmitter, delay (or skew) may be introduced into one or more of the channels based on characteristics of the transmitter, receiver, and/or the optical fiber. As a result, the relative timing of the data in the various channels may be misaligned at the receiver, causing degradation of the recovered data.

Although there are several types of devices and methods related to optical communication systems, they have been inadequate for the advancement of various applications. Conventional embodiments consume large areas or large amounts of power and suffer from performance limitations. Therefore, improved devices and methods for optical communication systems and related electronics are highly desired.

BRIEF SUMMARY OF THE INVENTION

The present invention relates to communication systems and integrated circuit (IC) devices. More particularly, the present invention provides for improved methods and devices for optical communication.

According to an example, the present invention provides a method and structure for an implementation of distribution matcher encoders and decoders for probabilistic shaping applications. The device can include a distribution matcher encoder module configured with the means to convert data from a first form to a second form according to an encoder method using probabilistic shaping. These methods can be implemented in hardware or in software. The device can also include a distribution matcher decoder module configured with the means to convert data from the second form to the first form according to a decoder method using probabilistic shaping. Similar to the encoder module, the decoder module can be implemented in hardware or in software. Further, a channel module electrically coupled between the distribution matcher encoder module and the distribution matcher decoder module. The distribution matcher encoder module can be provided on an egress path coupled to the channel module, while the distribution matcher decoder module is provided on an ingress path coupled to the channel module.

In a specific example, the egress path can include a splitter module electrically coupled to the distribution matcher encoder module and configured to receive data. The egress path can also include a forward error correction (FEC) encoder module electrically coupled to the splitter module and the distribution matcher encoder module. And, a mapper module can be electrically coupled to the FEC encoder module and the channel module. The ingress path can have a similar configuration with a demapper module electrically coupled to the channel module, a forward error correction (FEC) decoder module electrically coupled to the demapper module and the distribution matcher decoder module, and a merger module electrically coupled to the distribution matcher decoder module and the FEC decoder module, and configured to output data.

The methods and structures of the present invention provide many benefits for optical communication. Specifically, the techniques involved avoid the traditional implementations based on arithmetic coding, which requires intensive multiplication functions. Furthermore, these probabilistic shaping techniques can be used in combination with LDPC codes or any other Error Correction Code (such as Turbo Codes, Polar Codes, BCH/RS Codes, Braided Codes) through reverse concatenation techniques.

A further understanding of the nature and advantages of the invention may be realized by reference to the latter portions of the specification and attached drawings.

BRIEF DESCRIPTION OF THE DRAWINGS

In order to more fully understand the present invention, reference is made to the accompanying drawings. Understanding that these drawings are not to be considered limitations in the scope of the invention the presently described embodiments and the presently understood best mode of the invention are described with additional detail through the use of the accompanying drawings in which:

FIG. 1 is a simplified diagram illustrating an implementation architecture of a reverse concatenation scheme according to an example of the present invention.

FIG. 2 is a simplified diagram illustrating a distribution matcher encoder architecture according to an example of the present invention.

FIG. 3 is a simplified diagram illustrating a distribution matcher encoder function block according to an example of the present invention.

FIG. 4 is a simplified diagram illustrating a distribution matcher decoder architecture according to an example of the present invention.

FIG. 5 is a simplified diagram illustrating a distribution matcher decoder function block according to an example of the present invention.

FIG. 6 is a sample graph illustrating the spectral efficiency (SE) of probabilistic shaping using the distribution matcher implementation architectures according to examples of the present invention.

FIGS. 7A-7D are sample images of probabilistic shaped constellations using the distribution matcher implementation architectures according to examples of the present invention.

DETAILED DESCRIPTION OF THE INVENTION

The present invention relates to communication systems and integrated circuit (IC) devices. More particularly, the present invention provides for improved methods and devices for optical communication.

The following description is presented to enable one of ordinary skill in the art to make and use the invention and to incorporate it in the context of particular applications. Various modifications, as well as a variety of uses in different applications will be readily apparent to those skilled in the art, and the general principles defined herein may be applied to a wide range of embodiments. Thus, the present invention is not intended to be limited to the embodiments presented, but is to be accorded the widest scope consistent with the principles and novel features disclosed herein.

In the following detailed description, numerous specific details are set forth in order to provide a more thorough understanding of the present invention. However, it will be apparent to one skilled in the art that the present invention may be practiced without necessarily being limited to these specific details. In other instances, well-known structures and devices are shown in block diagram form, rather than in detail, in order to avoid obscuring the present invention.

The reader's attention is directed to all papers and documents which are filed concurrently with this specification and which are open to public inspection with this specification, and the contents of all such papers and documents are incorporated herein by reference. All the features disclosed in this specification, (including any accompanying claims, abstract, and drawings) may be replaced by alternative features serving the same, equivalent or similar purpose, unless expressly stated otherwise. Thus, unless expressly stated otherwise, each feature disclosed is one example only of a generic series of equivalent or similar features.

Furthermore, any element in a claim that does not explicitly state “means for” performing a specified function, or “step for” performing a specific function, is not to be interpreted as a “means” or “step” clause as specified in 35 U.S.C. Section 112, Paragraph 6. In particular, the use of “step of” or “act of” in the Claims herein is not intended to invoke the provisions of 35 U.S.C. 112, Paragraph 6.

Please note, if used, the labels left, right, front, back, top, bottom, forward, reverse, clockwise and counter clockwise have been used for convenience purposes only and are not intended to imply any particular fixed direction. Instead, they are used to reflect relative locations and/or directions between various portions of an object.

I. Overview

According to an example, the present invention provides a method and structure for an implementation of distribution matcher encoders and decoders for probabilistic shaping applications. The techniques involved avoid the traditional implementations based on arithmetic coding, which requires intensive multiplication functions. Furthermore, these probabilistic shaping techniques can be used in combination with LDPC codes or any other Error Correction Code (such as Turbo Codes, Polar Codes, BCH/RS Codes, Braided Codes) through reverse concatenation techniques.

In order to achieve Shannon capacity, constellation shaping must be applied. This can be achieved by using geometric shaping and/or probabilistic shaping. Here, we focus on probabilistic shaping because it can provide better flexibility than geometric shaping. However, conventional techniques for probabilistic shaping are based on the use of complex arithmetic coding technics, which require multipliers and high resolution arithmetic. According to an example, the present invention implements a novel technique based on LUTs and adders in order to achieve a high parallelism and high throughput architecture suitable for optical communications. The complexity of this architecture is moderate and easily implementable with the current technology. Furthermore, the proposed Distribution Matcher is optimal is the sense that it achieves the highest possible rate for a fixed block size.

In order to provide a practical implementation of this technique we combine probabilistic shaping with high coding gain LDPC code, which constitutes the state of the art in error correcting codes for optical communications. The combination is based on a well-known technique call reverse concatenation. Here, we describe an apparatus which is capable of producing numbers with the desired distribution either for cases of constant word weight, i.e., the desired probability is obtained in all codewords. On the other hand, another version is proposed in which the desired probability is obtained in the average of all the codewords. The latter scheme allows a higher coding efficiency.

Simulation results of a multi-level QAM modulation with probabilistic shaping are also provided. Also, an analysis of the pipelining of the forward and backward recursions present in the algorithm is carried out in order to provide some intuition about feasibility and obstacles in the practical implementation.

II. Probabilistic Shaping A. Distribution Matching (DM)

DMs transform a sequence of uniformly distributed input bits into an output sequence of symbols from an alphabet

with a desired distribution. A fixed-to-fixed length DM maps an input vector of k bits denoted d to an output vector of n symbols s=DM(d). The mapping DM is invertible, i.e., d can be recovered from a s by applying the inverse mapping DM⁻¹. The rate of the DM is as follows:

$\begin{matrix} {R_{DM} = \frac{k}{n}} & (1) \end{matrix}$ Let n_(α)(s) be the number of symbols α∈

present in the output vector s. If for each α∈

, the n_(α)(s) has the same value for all valid output vectors s, the DM is called Constant Composition Distribution Matcher (CCDM) and n_(α)(s) is denoted by n_(α). In this case, the input dimension k can be computed based on the multinomial distribution as follows:

$\begin{matrix} {k = \left\lfloor {\log_{2}\begin{pmatrix} n \\ {n_{a_{0}},n_{a_{1}},\ldots,n_{a_{q - 1}}} \end{pmatrix}} \right\rfloor} & (2) \end{matrix}$ where q=

; and α_(i)∈

for i=0, 1, . . . , q−1; and the output distribution is as follows:

$\begin{matrix} {{P_{A}(a)} = \frac{n_{a}}{n}} & (3) \end{matrix}$

An important metric of any particular implementation of the DM is its rate loss R_(loss). The rate loss is the difference between the entropy rate

(A) of a discrete memory-less source (DMS) with probability P_(A) and the DM rate

$\frac{k}{n},$ and is represented as follows:

$\begin{matrix} {R_{loss} = {(A) - \frac{k}{n}}} & (4) \end{matrix}$

The rate loss of CCDM vanishes for large output lengths n. In an example of the present invention, we are interested in DMs with relatively short output lengths and we therefore need to account for the rate loss in our system design.

In an example, the constraint imposed on n_(α)(s) by the CCDM must be avoided to reduce the rate loss. An example of an optimal DM algorithm without this constraint is also introduced in section III.

B. Modulation

We consider a constellation based on pulse amplitude modulation (PAM) with q=2^(m) points in each dimension. The set of these points is denoted by X, and is represented as follows: X={±1, ±3, . . . , ±(2^(m)−1)}  (5)

Let

denote the set of amplitudes of X, represented as follows:

={1, 3, . . . , (2^(m)−1)}={|x|:x∈X }  (6)

In an example, the present invention includes a label function β: X→{0, 1}^(m) and corresponding bits mappers _(x): {0, 1}^(m)→X. Let b=[b₁, b₁, . . . , b_(m)]∈{0, 1}^(m) . The label function β and the mapper _(x) are divided in two sub-functions with one for the amplitude and one for the sign of point x∈X, shown as follows:

$\begin{matrix} {x = \left\{ \begin{matrix} {- {x_{\mathcal{A}}\left( {b_{2},\ldots,b_{m}} \right)}} & {{{if}\ b_{1}} = 1} \\ {x_{\mathcal{A}}\left( {b_{2},\ldots,b_{m}} \right)} & {{{if}\ b_{1}} = 0} \end{matrix} \right.} & (7) \end{matrix}$ where _(X)

: {0, 1}^(m)→

maps bits to amplitudes and similarly its inverse function is denoted

→{0, 1}^(m−1).

The two classical implementations of

and

are based on the binary reflected Gray code (BRGC) and the natural based binary code (NBBC) where the amplitude label is a natural binary code (NBC).

B. Reverse Concatenation

FIG. 1 is a simplified diagram illustrating an implementation architecture of a reverse concatenation scheme according to an example of the present invention. As shown, device 100 includes an egress path 101 coupled to an ingress path 102 via a channel module 103. The egress path 101 receives data at a splitter module 111, which is electrically coupled to a distribution matching encoder module 121 and a forward error correction (FEC) encoder module 131, which is also electrically coupled to the distribution matching encoder module 121. In a specific example, the input data is split into random data and controlled data; the splitter module 111 sends the controlled data through the distribution matching encoder module 121 to the FEC encoder module 131 while sending the random data directly to the FEC encoder module 131. Also, the FEC encoder module 131 can include a parity module and a data module, and is also electrically coupled to a mapper module 141, which receives the data from the FEC encoder module 131 and outputs 2D symbols to the channel module 103. In a specific example, the FEC encoder module 131 sends the random data along with an output from the parity module to a sign input of the mapper module 141. Further, the FEC encoder module 131 sends the controlled data to an amplitude input of the mapper module 141.

The ingress path 102 reverses the process of the egress path 101. This path receives the 2D symbol data from the channel module 103 at a demapper module 142, which is electrically coupled to an FEC decoder module 132. In a specific example, the demapper module 142 sends the encoded random data and parity data from a sign output, and also sends the encoded controlled data from an amplitude output. Similar to the FEC encoder module 131, the FEC decoder module 132 can include a parity module, which receives the parity data, and a data module, which receives the random and controlled data. The FEC decoder module 132 is electrically coupled to a distribution matching decoder module 122 and a merger module 112. In a specific example, the FEC decoder module 132 sends the random data directly to the merger module 112 while sending the controlled data through the distribution matching decoder 122 to the merger module 112. The merger module 112 then outputs the decoded and merged data.

According to an example, the DM and the FEC code are concatenated in the reverse order as described in FIG. 1 in order to simplify the implementation of the DM decoder and to avoid error propagation. In this example, the DM maps k data bits to n amplitudes, which are represented by a vector of n(m−1) bits. Consider that the FEC code has {tilde over (k)} dimension and ñ length. Then several blocks of these amplitude bits, referred to as controlled bits in FIG. 1 , are grouped in a block of αn(m−1) bits. These bits and αγn additional data bits, called random bits, form a block of k=αn(m−1)+αγn, which has to be multiplied with the parity generating part P of a systematic generator matrix [I|P] of the FEC to generate α(1−γ)n redundancy bits. The redundancy bits and the additional data bits are mapped to an signs, which are multiplied symbol-wise with the αn amplitudes. The FEC code instantiated by P has a rate determined as follows:

$\begin{matrix} \begin{matrix} {c = \frac{\overset{\sim}{k}}{\overset{\sim}{n}}} \\ {c = \frac{{a{n\left( {m - 1} \right)}} + {a\gamma n}}{amn}} \\ {c = {\frac{{n\left( {m - 1} \right)} + {\gamma n}}{mn} = \frac{m - 1 + \gamma}{m}}} \end{matrix} & (8) \end{matrix}$

The fraction of signs used for data bits is as follows: γ=1−(1−c)m   (9)

The reverse concatenation scheme requires 0≤γ≤1. The transmission rate of the reverse scheme is the number of data bits per PAM symbol given by the following:

$\begin{matrix} {R_{t} = {\frac{k}{n} + \gamma}} & (10) \end{matrix}$

From the point of view of efficiency, CCDM is not the most optimum option since it requires the distribution to be satisfied between all vectors in the codebook for the R_(loss) to be reduced. A minor modification is discussed later in order to address this aspect.

III. Distribution Matcher Algorithm

The majority of distribution matcher implementations are based on arithmetic coding techniques. These techniques are very efficient for data compression where variable length coding is desirable. However, for constant composition distribution matcher (CCDM) where fixed length coding is preferable, arithmetic coding is not straightforward. Furthermore, arithmetic coding requires multiplication operations in each step, which significantly increase its implementation complexity.

In an example, the present invention includes an implementation of a new DM algorithm using a small number of addition and comparison operations and completely avoiding multiplication or division operations. This scheme is also suitable for parallel implementation, as required in high speed signal processing. In addition, the proposed algorithm is also flexible and can easily be configured to provide different target distributions with the same base architecture. In fact, by using either programmable lookup tables (LUTs) or fixed LUTs designed for a desired maximum output weight, the same hardware will work for smaller weights.

The following presents six examples of proposed algorithms for binary output alphabets that can be implemented in various examples of the present invention. The implementations can include both software and hardware types. These algorithms can also be generalized to higher order alphabets in a straightforward manner. Those of ordinary skill in the art will recognize other variations, modifications, and alternatives.

According to an example, the present invention incorporates an encoder algorithm, shown below in Algorithm 1. Here, the input parameters are the code length n, the number of bits per codeword n′, and the data input x.

Algorithm 1—Algorithm for Distribution Matcher Encoder I

-   Input: x, n′, n -   Output: b     -   1: α_(n)=x     -   2: n′_(n =n′)     -   3: for i=n−1 to 0 do     -   4: if

$\left( {a_{i + 1} \geq \begin{pmatrix} i \\ {n_{i + 1}^{\prime} - 1} \end{pmatrix}} \right.$ or n′_(i+1)≤0) then

-   -   5:

$a_{i} = {a_{i + 1} - \begin{pmatrix} i \\ {n_{i + 1}^{\prime} - 1} \end{pmatrix}}$

-   -   6: b_(i)=0     -   7: n′_(i)=n′_(i+1)     -   8: else     -   9: α_(i)=α_(i+1)     -   10: b_(i)=1     -   11: n′_(i)=n′₁₊₁−1     -   12: end if     -   13: end for     -   14: return b

According to an example, the present invention incorporates a decoder algorithm, shown below in Algorithm 2. This decoder algorithm corresponds to the encoder algorithm of Algorithm 1.

Algorithm 2—Algorithm for Distribution Matcher Decoder I

-   Input: b, n′, n -   Output: x     -   1: α_(n)=0     -   2: n′_(n)=n′     -   3: for i=n−1 to 0 do     -   4: if (b_(i)==1 or n′_(i+1)==0) then     -   5: α_(i)=α_(i+1)     -   6: else     -   7:

$a_{i} = {a_{i + 1} + \begin{pmatrix} i \\ {n_{i + 1}^{\prime} - 1} \end{pmatrix}}$

-   -   8: end if     -   9: if (b_(i)==1) then     -   10: n′_(i)=n′_(i+1)−1     -   11: else     -   12: n′_(i)=n′_(i+1)     -   13: end if     -   14: end for     -   15: x=α₀     -   16: return x

Algorithms 3 and 4 (below) are different versions of Algorithms 1 and 2, respectively, based on a LUT approach. The implementation architecture of Algorithm 3 is further described in FIGS. 2 and 3 , while that of Algorithm 4 is described in FIGS. 4 and 5 .

Algorithm 3—Algorithm for Distribution Matcher Encoder I with LUT

-   Input: x, n′, n -   Output: b     -   1: α_(n)=x     -   2: n′_(n)=n′     -   3: for i=n−1 to 0 do     -   4: if (α_(i+1)≥t_(i)[n′_(i+1)−1] or n′_(i+1)≤0) then     -   5: α_(i)=α_(i+1)−t_(i)[n′_(i+1)−1]     -   6: b_(i)=0     -   7: n′_(i)=n′_(i+1)     -   8: else     -   9: α_(i)=α_(i+1)     -   10: b_(i)=1     -   11: n′_(i)=n′_(i+1)−1     -   12: end if     -   13: end for     -   14: return b

According to an example, the present invention uses carry save adders to implement these algorithms in high-speed optical systems. In these cases, two blocks are required to implement a fully operable system with probabilistic shaping: a distribution matcher encoder, and a distribution matcher decoder.

FIG. 2 is a simplified diagram illustrating a distribution matcher encoder architecture according to an example of the present invention. As shown, device 200 includes a series of n function blocks coupled together in series. Here, a first function block 211 denoted “Λ_(n−1),” a second function block 212 denoted “Λ_(n−2),” and a final function block 219 denoted “Λ₀” separated by an ellipsis represent the connection between the n function blocks. Further of the function block are provided in FIG. 3 .

FIG. 3 is a simplified diagram illustrating a distribution matcher encoder function block according to an example of the present invention. More specifically, this device 300 is an example unit architecture configured to perform Algorithm 3 using an LUT approach and addition/subtraction computations. As shown, device 300 receives inputs x (denoted “i_x”) and n′ (denoted “i_n1”) and includes adders 311 and 312; an LUT multiplexer 320; an AND gate (with an inverted input) 330; multiplexers 341 and 342; and storage units 351 (outputting “o_n1”), 352 (outputting “o_x”), and 353 (outputting “o_bit”). Those of ordinary skill in the art will recognize other variations, modifications, and alternatives to implementations of the encoder algorithms discussed herein.

Algorithm 4—Algorithm for Distribution Matcher Decoder I with LUT

-   Input: b, n′, n -   Output: x     -   1: α_(n)=0     -   2: n′_(n)=n′     -   3: for i=n−1 to 0 do     -   4: if (b_(i)==1 or n′_(i+1)==0) then     -   5: α_(i)=α_(i+1)     -   6: else     -   7: α_(i)=α_(i+1)+t_(i)[n′_(i+1)−1]     -   8: end if     -   9: if (b_(i)==1) then     -   10: n′_(i)=n′_(i+1)−1     -   11: else     -   12: n′_(i)=n′_(i+1)     -   13: end if     -   14: end for     -   15: x=α₀     -   16: return x

From comparing algorithm 1 and algorithm 3, the following relation for the LUT values can be deduced as follows:

${t_{i}\left\lbrack {n_{i + 1}^{\prime} - 1} \right\rbrack} = {\begin{pmatrix} i \\ {n_{i + 1}^{\prime} - 1} \end{pmatrix}.}$ But the LUT values can be smaller than these values and the encoder and decoder algorithms will still work, with the loss of some valid words of the encoder output alphabet. This may not represent a loss of performance in the CCDM due to the flooring operation in equation (2). Modifying the LUT values gives flexibility to the design of the blocks and can be used for several purposes; for instance, to save power.

FIG. 4 is a simplified diagram illustrating a distribution matcher decoder architecture according to an example of the present invention. As shown, device 400 includes a series of n function blocks coupled together in series. Here, a first function block 411 denoted “Λ⁻¹ _(n−1), ” a second function block 412 denoted “Λ⁻¹ _(n−2),” and a final function block 419 denoted “Λ⁻¹ ₀” separated by an ellipsis represent the connection between the n function blocks. These function blocks correspond to the encoder function blocks shown in FIG. 2 . Further of the function block are provided in FIG. 5 .

FIG. 5 is a simplified diagram illustrating a distribution matcher decoder function block according to an example of the present invention. More specifically, this device 500 is an example unit architecture configured to perform Algorithm 4 using an LUT approach and addition/subtraction computations. As shown, device 500 receives inputs b (denoted “i_bit”), x (denoted “i_x”) and n′ (denoted “i_n1”) and includes adders 511 and 512; an LUT multiplexer 520; an OR gate 530; multiplexers 541 and 542; and storage units 551 (outputting “o_n1”) and 552 (outputting “o_x”). Those of ordinary skill in the art will recognize other variations, modifications, and alternatives to implementations of the decoder algorithms discussed herein.

In order to reduce R_(loss), the present invention can also incorporate modified versions of algorithms 1, 2, 3, and 4, which are shown below in Algorithms 5 and 6. According to a specific example, these modified algorithms avoid the constant compression constraint in n_(α)(s) and achieves the minimum possible R_(loss) for a given codeword length n.

Algorithm 5—Algorithm for Distribution Matcher Encoder II

-   Input: x, n -   Output: b     -   1: n′=0     -   2: while

$\left( {x \geq \ \begin{pmatrix} n \\ n^{\prime} \end{pmatrix}} \right)$ do

-   -   3:

$x = {x - \begin{pmatrix} n \\ n^{\prime} \end{pmatrix}}$

-   -   4: n′=n′+1     -   5: end while     -   6: α_(n)=x     -   7: for i=n−1 to 0 do     -   8: if (α_(i+1)≥t_(i)[n′−1] or n′≤0) then     -   9: α_(i)=α_(i+1)−t_(i)[n′−1]     -   10: b_(i)=0     -   11: else     -   12: α_(i)=α_(i+1)     -   13: b_(i)=1     -   14: n′=n′−1     -   15: end if     -   16: end for     -   17: return b

For the modified encoder, the implementation architecture is equivalent to that described in FIGS. 2 and 3 with the additional sub-algorithm shown in lines 1 to 6 above in Algorithm 5. Those of ordinary skill in the art will recognize other variations, modifications, and alternatives.

Algorithm 6—Algorithm for Distribution Matcher Decoder II

-   Input: b, n -   Output: x     -   1: n′_(n)ω_(H)(b)     -   2:

$a_{n} = {\sum\limits_{k = 1}^{\omega_{H}(b)}\begin{pmatrix} n \\ k \end{pmatrix}}$

-   -   3: for i=n−to 0 do     -   4: if (b_(i)==1 or n′_(i+1)==0) then     -   5: a_(i)=α_(i+1)     -   6: else     -   7: α_(i)=α_(i+1)+t_(i)[n′_(i+1)−1]     -   8: end if     -   9: if (b_(i)==1) then     -   10: n′_(i)=n′_(i+1−)1     -   11: else     -   12: n′_(i)=n′_(i+1)     -   13: end if     -   14: end for     -   15: return x

For the modified decoder, the implementation architecture is equivalent to the architecture described in FIGS. 4 and 5 with the additional sub-algorithm described in lines 1 to 2 of Algorithm 6. Here, the function ω_(H)(b) is the Hamming weight of b (i.e., the number of elements with a value different from zero). The α_(n)function is a summation equation of the

$\begin{pmatrix} n \\ k \end{pmatrix}$ term from k=1 to k=ω_(H)(b). Of course, there can be other variations, modifications, and alternatives.

In an example the present invention provides for a method and structure of a communication device. The device can include a distribution matcher encoder module configured with the means to convert data from a first form (x) to a second form (b) according to an encoder method, such as Algorithms 1, 3, and 5 discussed previously. These methods can be implemented in hardware, such as the examples shown in FIGS. 2 and 3 , or in software, which can involve a code programmed with an encoder method that is stored in a memory unit and performed by a processor.

The device can also include a distribution matcher decoder module configured with the means to convert data from the second form to the first form according to a decoder method, such as Algorithms 2, 4, and 6 discussed previously. Similar to the encoder module, the decoder module can be implemented in hardware, such as the examples shown in FIGS. 4 and 5 , or in software, which can involve a code programmed with a decoder method that is stored in a memory unit and performed by a processor.

Further, a channel module electrically coupled between the distribution matcher encoder module and the distribution matcher decoder module. In a specific example, the configuration can be similar to that shown in FIG. 1 , having an egress path of modules feeding the input data to be encoded and sent to the channel module, which sends the encoded data through the ingress path to be decoded and outputted. Those of ordinary skill in the art will recognize other variations, modifications, and alternatives.

IV. Distribution Matcher Simulation Results

FIGS. 6 and 7A-7D show simulation results from test implementations of the proposed distribution matcher algorithms in combination with an LDPC code according to an example of the present invention.

FIG. 6 is a sample graph illustrating the spectral efficiency (SE) of probabilistic shaping using the distribution matcher implementation architectures according to examples of the present invention. As shown, graph 600 shows simulation results of data rate over required SNR for TDHQ, 64-QAM, 32-QAM, 16-QAM, 8-QAM, and QPSK.

FIGS. 7A-7D are sample images of probabilistic shaped constellations using the distribution matcher implementation architectures according to examples of the present invention. FIG. 7A shows image 701 representing a signal constellation of an 8-QAM (quadrature amplitude modulation) form with SE=2.78 bits/symbol. FIG. 7B shows image 702 representing a signal constellation of a 16-QAM from with SE=3.78 bits/symbol. FIG. 7C shows image 703 representing a signal constellation of a 32-QAM form with SE=4.78 bits/symbol. And FIG. 7D shows image 704 representing a signal constellation of a 64-QAM form with SE=5.67 bits/symbol.

While the above is a full description of the specific embodiments, various modifications, alternative constructions and equivalents may be used. Therefore, the above description and illustrations should not be taken as limiting the scope of the present invention which is defined by the appended claims. 

What is claimed is:
 1. A transceiver comprising: a splitter configured to receive a first sequence of bits of unknown composition and split the first sequence of bits into a second sequence of bits and a third sequence of bits, wherein bits of the first sequence of bits are random and have an equal probability of occurring in the first sequence of bits; a distribution matching encoder configured to generate a fourth sequence of bits based on the third sequence of bits such that the fourth sequence of bits has a restricted number of ‘1’ bits and a restricted number of ‘0’ bits per codeword of the fourth sequence of bits, wherein the fourth sequence of bits includes a greater quantity of total bits than the third sequence of bits, and wherein bits of the fourth sequence of bits have controlled probabilities of occurring in the fourth sequence of bits and do not have equal probabilities of occurring in the fourth sequence of bits; and a mapper configured to generate symbols having respective signs and amplitudes, set the signs of the symbols based on the second sequence of bits, set the amplitudes of the symbols based on the fourth sequence of bits, and transmit the symbols from the transceiver.
 2. The transceiver of claim 1, wherein the distribution matching encoder is configured to generate the fourth sequence of bits including a plurality of codewords such that each of the plurality of codewords includes a same number of ‘1’ bits and a same number of ‘0’ bits.
 3. The transceiver of claim 1, wherein the distribution matching encoder is configured to generate the fourth sequence of bits including a plurality of codewords such that each of the plurality of codewords has a same average transmission energy after the fourth sequence of bits are converted to symbols by the mapper.
 4. The transceiver of claim 1, wherein the mapper is configured to generate the symbols having a two-dimensional signal constellation with a quadrature amplitude modulation form.
 5. The transceiver of claim 1, wherein the distribution matching encoder is configured to generate the fourth sequence of bits using addition and comparison operations and without using multiplication and division operations.
 6. The transceiver of claim 1, further comprising a forward error correction encoder configured to add parity bits to the second sequence of bits prior to the second sequence of bits being received at the mapper.
 7. The transceiver of claim 1, wherein the distribution matching encoder is configured to transform the third sequence of bits, which have a uniformly distributed constellation, to the fourth sequence of bits having a non-uniformly distributed constellation.
 8. The transceiver of claim 1, wherein the distribution matching encoder is configured to generate the fourth sequence of bits such that bits of the fourth sequence of bits have controlled probabilities of occurring, where ‘1’ bits of the fourth sequence of bits have less probability of occurring than ‘0’ bits of the fourth sequence of bits.
 9. The transceiver of claim 1, wherein the distribution matching encoder is configured to implement an iterative algorithm to output the fourth sequence of bits including performing a plurality of comparisons to bit values of the third sequence of bits, the plurality of comparisons are performed based on a code length of the third sequence of bits and a number of bits per codeword of the fourth sequence of bits.
 10. The transceiver of claim 1, wherein the distribution matching encoder is configured to implement an iterative algorithm to output the fourth sequence of bits including performing a plurality of comparisons between bit values of the third sequence of bits and look-up-table values.
 11. A transceiver comprising: a demapper configured to receive symbols transmitted to the transceiver, determine signs and amplitudes of the symbols, generate a first sequence of bits based on the signs, and generate a second sequence of bits based on the amplitudes, wherein bits of the first sequence of bits have an equal probability of occurring, bits of the second sequence of bits have controlled probabilities of occurring and do not have equal probabilities of occurring, and the second sequence of bits has a restricted number of ‘1’ bits and a restricted number of ‘0’ bits per codeword received; a distribution matching decoder configured to generate a third sequence of bits based on the second sequence of bits, wherein bits of the third sequence of bits have an equal probability of occurring in the third sequence of bits, and wherein the third sequence of bits includes fewer total bits than the second sequence of bits; and a merging device configured to merge the first sequence of bits and the third sequence of bits to provide a resulting random sequence of bits.
 12. The transceiver of claim 11, wherein the demapper is configured to generate the second sequence of bits including a plurality of codewords such that each of the plurality of codewords includes a same number of ‘1’ bits and a same number of ‘0’ bits.
 13. The transceiver of claim 11, wherein the symbols have a two-dimensional signal constellation with a quadrature amplitude modulation form.
 14. The transceiver of claim 11, wherein the distribution matching decoder is configured to generate the third sequence of bits using addition and comparison operations and without using multiplication and division operations.
 15. The transceiver of claim 11, further comprising a forward error correction decoder configured to remove parity bits from the first sequence of bits prior to the first sequence bits being received at the merging device.
 16. The transceiver of claim 11, wherein the distribution matching decoder is configured to transform the second sequence of bits, the second sequence of bits having a non-uniformly distributed constellation, to the third sequence of bits having a uniformly distributed constellation.
 17. The transceiver of claim 11, wherein the demapper is configured to generate the second sequence of bits such that bits of the second sequence of bits have controlled probabilities of occurring, where ‘1’ bits of the second sequence of bits have less probability of occurring than ‘0’ bits of the second sequence of bits.
 18. The transceiver of claim 11, wherein the distribution matching decoder is configured to implement an iterative algorithm to output the third sequence of bits including performing a plurality of comparisons to bit values of the second sequence of bits, where the plurality of comparisons are performed based on a code length of the third sequence of bits and a number of bits per codeword of the second sequence of bits.
 19. The transceiver of claim 11, wherein the distribution matching decoder is configured to implement an iterative algorithm to output the third sequence of bits including performing a plurality of comparisons between bit values of the second sequence of bits and look-up-table values. 